Smith-Kettlewell TECHNICAL FILE

Published by Rehabilitation Engineering Center, Smith-Kettlewell Institute of Visual Sciences

Bill Gerrey, Editor

Supported, in part, by Smith-Kettlewell Eye Research Foundation and National Institute of Handicapped Research

Produced by

[Spring 1983, Vol. 4, No. 2]

Table of contents

Operational Amplifiers II

The National LM317 and LM337 Adjustable Voltage Regulators

National Semiconductor Stereo Power Amplifier IC's

The Editor's Evaluation of the DRI Industries' One-Hand Soldering Iron

The JA3TBW Solder Guide

Operational Amplifiers II

By Albert Alden

Introduction

Last issue in the first article in this series we described the ideal errorless op-amp and gave some applications. This issue we will treat the limitations of practical op-amps and show a number of additional applications.

We will first discuss the static errors and then the dynamic errors.

Static Characteristics

Input Offset Voltage

With an ideal op-amp, shorting the input terminals together and connecting them to ground results in an output of zero volts. In a real op-amp, a small voltage must be applied between the input terminals to obtain zero volts out. This externally applied voltage cancels an internal offset voltage error of the opposite polarity. The source of this error is a mismatch between the base-emitter voltages of the input transistors of the inverting and non-inverting inputs. Eos is typically 2 millivolts for a 741. For both the inverting and non-inverting amplifier circuits, the error due to E offset is given by

Eo = Eos (1 + (Ro/Ri))

or the error is equal to the quantity one plus the gain all times the offset, for an inverting circuit; and for the non-inverting circuit, the gain times the offset.

For example, an inverting amplifier with a gain of 10 using an op-amp with an input offset voltage of 2 millivolts will have an output error of 22 millivolts.

Note that the input offset voltage specified may be of either polarity.

In many op-amps the offset voltage may be nulled with an external potentiometer. For the 741, a 10K pot is connected between pins 1 and 5, and the wiper is connected to the negative power supply (-V). What this does is give one side or the other of the differential input stage more or less current, thus causing a base-emitter voltage change to cancel the initial mismatch.

Precision op-amps with low offset voltages are available which can be used where trimming isn't desired.

If in your application you don't need gain at DC, then capacitively coupling the output eliminates the offset voltage error problem.

Offset voltage changes with temperature; a change of 5uV per degree centigrade is typical for general purpose op-amps such as the 741.

Input Bias and Input Offset Current

All op-amps require that a certain DC current be supplied at each input. This current ranges from picoAmps to microAmps, depending on the device. This is called the input bias current (Ib). The value given in specifications for op-amps is the mean of the currents for the inverting and non-inverting inputs. The 741 has a typical bias current of 80 nanoAmps. (Note: Don't confuse this parameter with input impedance; they are not the same.)

The error due to Ib for both the inverting and non-inverting amplifier circuits is given by Eo = IbRo, where Ib is the input bias at the inverting input of the op-amp.

For example, with Ib equals 80 nanoAmps and Ro equals 100K, the output error will be equal to 8 millivolts.

This error can be reduced by taking advantage of the bias current into the non-inverting input of the op-amp. By placing a resistor in series with the connection to the non-inverting input (i.e., between the non-inverting input and ground for the inverting amplifier circuit, or between the non-inverting input and the signal source for the non-inverting amplifier), the output error then becomes a function of the difference in the two bias currents. This difference is called the input offset current, Ios. The value of the series resistor is selected to be equal to Ro in parallel with Ri.

R3 = (RoRi/(Ro + Ri)

The output error is then Eo = IosRo.

For the 741, Ios is typically 20 nanoAmps. This will give an error of 2 millivolts in the above example with the proper value of R3 inserted in the circuit.

For each application the user should check to see if the offset is low enough without R3. If too high, calculate the offset with R3 to see if it meets your requirements. If not, consider using lower value resistors in your circuit or an FET input op-amp where Ib is typically 20 picoAmps.

Input bias and input offset current change with temperature. Typical values for the 741 are:

Input bias current--0.25 nanoAmps per deg. C.
Input offset current--0.025 nanoAmps per deg. C.

Gain

The gain of our ideal op-amp was assumed to be infinite. In actuality it is quite high, ranging typically from 100,000 to one million. In feedback terminology, this is called the open loop gain.

We will digress a bit at this point to say a few words about feedback systems in order to understand the effect of gain on op-amp circuits.

A feedback system is one in which some function or portion of the output is fed back to, and combined with, the input to the system. In a negative feedback system, the portion of the output fed back is subtracted from the input of the system before being amplified. Positive feedback systems add the feedback to the input. Positive feedback usually results in oscillation.

In general, negative feedback does everything good for a system--it increases bandwidth, improves linearity, reduces distortion, reduces output impedance, increases input impedance, and stabilizes the gain. The main detrimental effect of negative feedback is a reduction in the gain of a system.

The gain of an amplifier with negative feedback is G equals A divided by the quantity l plus A times B, where

G is the closed loop gain (Eo over Ei)
A is the open loop or forward gain
B is the fraction of the output fed back

This equation can be applied to op-amp circuits.

For our non-inverting amplifier circuit, B = Ri/(Ro + Ri) (i.e., the input and output resistors form an attenuator).

Note that if the product A times B is much larger than one, the equation for G simplifies to G = l/B, which becomes

G = Ei((Ro/Ri) +1)

which is the same expression derived in last quarter's article.

If we put in some actual values in the equation for Eo/Ei, we can see the effect of not having infinite gain in the op-amp.

Let: A = 200,000
Ri = 1K
Ro = 9K
Then: B equals 0.l

We get G equals 9.9995--not much of an error!

The quantity AB is called the loop gain of the system and can be considered a figure of merit. It is approximately the factor by which the open loop gain is reduced and its reciprocal (l/AB) is approximately the error made in assuming infinite gain. In the above example, l/AB = 0.005%.

Analyzing the inverting amplifier op-amp circuit as a feedback system is not as obvious as for the non-inverting amplifier circuit. Skipping the proof we get:

A' = (Ro/(Ro + Ri))A
B = Ri/Ro
Loop gain = ARi/(Ro + Ri)

A' is used to indicate that the forward gain in this case is not the same as the open loop gain A. [Editor's Note: "A" followed by an apostrophe is standard mathematical notation; it is pronounced "A prime."]

End of digression. Conclusion: Infinite gain is a pretty good assumption.

Input and Output Impedance

Although the input impedance of an op-amp is not infinite and the output impedance is not zero, these parameters may be considered to be such in op-amp applications using negative feedback. They are each improved by a factor approximately equal to the loop gain.

Dynamic Characteristics

In any multistage amplifier such as an op-amp, each stage has a finite frequency response which is usually determined by its output resistance and a capacitance. This R and C form a low-pass filter resulting in a rolloff of 6dB per octave above the cutoff frequency. Associated with the rolloff is a phase lag of the signal approaching 90dg, or one quarter period. As the rolloff for each section comes into play at higher frequencies, the open loop gain rolls off faster at an additional 6dB per octave and an eventual 90dg lag for each section. This presents a problem.

At some frequency the open loop phase lag will reach 180dg or, in other words, a sign change. If we were to apply feedback around the op-amp, we would have positive feedback; and if the loop gain were greater than one at the 180dg phase lag frequency, we would have an oscillator. To make an op-amp a versatile device which can be used in circuits with wide ranges of closed loop gains, we must assure an open loop phase lag of less than 180dg for all frequencies where the open loop gain is greater than one.

This is done by adding enough capacitance to that stage of the op-amp circuit with the lowest frequency rolloff so that the open loop gain drops to below one at the frequency of the start of the rolloff of the second lowest frequency section. We now have an op-amp which has 90dg (or less) phase lag up to the frequency where the gain has rolled off to one. This scheme is called dominant pole compensation. This gives us a device which is stable for a wide range of closed loop gains. The unity gain frequency is called Ft. For the 741, Ft is typically 1.2 megaHertz.

The compensation capacitor comes built-in for most op-amps, including the 741. Others, called uncompensated op-amps, require an external capacitor, usually around 30 picoFarads. These units are versatile in that different compensations may be used to increase or decrease bandwidth to match particular applications. The uncompensated 74l is a 748; the external capacitor goes between pins l and 8.

With the open loop gain falling off with increase in frequency, there will be some point where the loop gain is not sufficient to keep the closed loop gain (G) dependent only on Ri and Ro. For dominant pole compensation op-amps, the closed loop corner frequency is given by the following equations:

Non-inverting amplifier: Fc = Ft/G
Inverting amplifier: Fc = Ft/(G + 1)

where Ft is the unity gain frequency. At Fc, the gain is down 3dB, or 0.707 of the low-frequency value, and the output signal has picked up a 45dg lag. Above Fc, the gain falls off at an initial rate approaching 6dB per octave.

An example: What will Fc be for a non-inverting amplifier with a gain of 100 using a 741?

Fc = 1.2 megaHertz/100 = 12,000 Hertz

Not too good for hi-fi systems.

Two solutions:

  1. Use a faster op-amp--one with Ft greater than 2 megaHertz.
  2. Divide the gain between two amplifiers, each with a gain of 10.

Using a 741 in the second solution, each section will have a bandwidth of 120,000 Hertz and a combined bandwidth (down 3dB) of 64% of 120,000, or 76,800 Hertz.

Input and Output Impedance vs. Frequency

It should be noted that the input and output impedances change with frequency. The input impedance of the 741 drops to about 400K ohms at one megaHertz, and the output impedance goes up to about 300 ohms at one megaHertz. Also, since the open loop gain is falling off with increase in frequency, the effect of these parameters on closed loop performance cannot be neglected, as was the case at DC.

Slew Rate Limit

The slew rate limit of an op-amp is the maximum rate at which the output voltage can change. This limit is caused because the internal compensation capacitor has available only finite currents for charging or discharging. For the 741, the limit is 0.5 volts per microsecond. If a signal is applied to an amplifier circuit at an amplitude and frequency requiring the output to change faster than the limit, the output waveform becomes distorted. A slew rate limited sinewave takes on a triangular shape.

The maximum rate of change for a sinewave occurs at the zero crossing and is a function of frequency and peak amplitude.

The maximum rate of change is given by the formula

Sr = 2π FVp

(or Sr equals 2 times pi times F times Vp),

where Sr is the maximum slew rate (which occurs at zero crossing).

F is the frequency of the sinewave
Vp is the peak voltage of the sinewave

By rearranging the above formula, we can calculate the maximum undistorted frequency of a sinewave output of amplitude Vp using an op-amp with a slew rate limit of Sr.

F = Sr/(2πV)p

(or F equals Sr divided by the product of 2 times pi times Vp)

For our 741 and Vp of 4 volts, F equals 20,000 Hertz--marginal for hi-fi applications.

A Better Op-Amp

Some of the limitations of the 74l have become obvious during the discussion of op-amp errors. While there are numerous op-amps of superior performance to the 74l, we will mention just one of them in this installment. Next issue we will list several and their important features.

The Texas Instruments TL07l

This unit is my present choice as an all-round, inexpensive op-amp for the vast majority of applications. It is an internally compensated FET input device with standard (74l) pin connections. The recommended input offset voltage-nulling circuit is slightly different from the 74l. A l00K pot is connected between pins l and 5, with the wiper going through l.5K ohms to the negative supply.

Here is a summary of specifications of the TL07l. The values given are typical.

Eos---3 millivolts
Ib--------30 picoAmps
Ios---5 picoAmps
A------------200,000
Ft---------3 megaHertz
Sr--------l3 volts per microsecond

Some More Op-Amp Circuits

Here are descriptions of three circuits using an op-amp and a diode to simulate an ideal diode. Power supply values are assumed to be the standard plus and minus l5V.

Active Peak Detector

The input to the circuit is the non-inverting input of an op-amp. Connected to the output of the op-amp is the anode of a diode, a lN4l48, for example. The cathode is connected to a capacitor whose other lead is grounded. The diode-capacitor junction is connected to the inverting input terminal of the op-amp. Also, a connection is made from this point to the non-inverting input of a second op-amp. The output of this op-amp is connected to its inverting input; i.e., it is connected as a follower. The output of the second op-amp is the output of the circuit.

Here is an application where an FET input op-amp with its low input current is required to minimize charge or discharge of the capacitor when holding the peak signal. The choice of capacitor value is a compromise between "droop" error and the ability of the op-amp to supply the current required for changing the capacitor's charge quickly enough to follow the input signal.

Absolute Value Circuit

This circuit appeared in the November 25, l982 issue of Electronic Design and was submitted to that magazine by Stan Rubin of Ragen Data Systems.

The output of the circuit is a positive voltage equal to the absolute value of the input signal.

The input of the circuit goes through a 50K ohm l% resistor to the inverting input of the first (of two) op-amps. The output of this op-amp is connected to the anode of a diode (lN4l48) whose cathode goes to the non-inverting input of the second op-amp. From the non-inverting input of the second op-amp, there is l0K going back to the circuit's input. The output of the second op-amp is connected to its inverting input, and there is a 50K l% resistor going from the output of the second op-amp to the inverting input of the first op-amp. The circuit's output is the output of the second op-amp.

Precision Clamp

This circuit consists, in its simplest form, of one resistor, one diode, and one op-amp. The input is on one side of a resistor (l0K is a good value). The other side of the resistor is the output, and it is connected to the inverting input of an op-amp. For positive limiting, the diode is connected from the inverting input to the output of the op-amp, with the anode at the input. To limit in the negative direction, turn the diode around.

The non-inverting input is connected to the voltage level we wish the signal to limit at. This may be a Zener diode, a voltage regulator, a resistor divider across a regulated supply, etc. A voltage follower on the output is needed if a low output impedance is required.

In the three above circuits, the speed is limited by the time needed for the op-amp to recover from saturation and the slew rate limit.

Bandwidth Control on the Inverting Amplifier

It is often desirable to limit the bandwidth of an amplifier more so than the limit imposed by the op-amp. A single-pole, low-frequency and high-frequency limit can be implemented, each with a capacitor. The low-frequency limit is set with a capacitor in series with the input resistor, Ri, and the high-frequency limit by a capacitor in parallel with Ro.

In both cases, the equation for the limiting frequency is

F = l/(2πRC)

(or F = 1 divided by the product of 2 times pi times R times C)

This is the frequency where the response (or gain) is 0.707 of the bandpass value (the -3dB point).

(Note: Cascading a pair of identical amplifiers will raise the lower limit frequency by l.56 and lower the upper frequency limit by 0.64.)

References

A great deal of the information contained in this article is covered in the Application Notes published by National Semiconductor, available together in the Linear Application Handbook.

Also, I highly recommend the following book as a very readable text on electronics; it is excellent.

Horowitz, P. and Hill, W.
The Art of Electronics
Cambridge University Press, l980.

Next month: A few more words about op-amp errors, a "Chipography," and some more applications.

The National LM317 And LM337 Adjustable Voltage Regulators

Abstract

The LM3l7 (positive) and LM337 (negative) voltage regulators lend themselves to "bootstrapping" in order to make adjustable power supplies. Various circuit arrangements are described, including a high-current supply using pass transistors, and a high-voltage supply using the LM3l7 in conjunction with a triode vacuum tube.

Theory of Operation

[Editor's Note: Don't give up on these devices if you do not immediately understand this section; this brief ? explanation is geared to the student of op-amps. The circuits described won't know if you've read this part or not.]

These units are basically high-current op-amps in the voltage-follower connection. They maintain a constant output voltage by comparing the IR drop across a "sensing resistor" in the boot-strapping circuit with an internal voltage standard. Since the positive and negative units (LM3l7 and LM337, respectively) are complementary, analysis need only be done on one of them, the 3l7.

These animals are very similar to their earlier counterparts, the "Three-Terminal Monolithic Voltage Regulators" discussed in SKTF, Fall l980. There are two important differences:

  1. Their "Adjustment" pin, analogous to the "common" terminal on their earlier counterparts, introduces very little current of its own into a boot-strapping circuit--typically 50uA as compared to 5mA of the older units.
  2. They are all 1.25 volt regulators which are intended for bootstrapping, their basic function being that of "series-pass regulators" for use at voltages above this value.

Their connection is as follows: The "Input" terminal goes to an unregulated voltage source. The "Output" terminal is the hot output of the supply, with the cold output of the supply being ground. This "Output" also goes through a "sensing resistor," R1 (perhaps 240 ohms), to the "Adjust" pin; this "Adjust" pin then goes through a rheostat, R2 (perhaps 5K), to ground.

The internal circuitry of the device (which is complicated past all understanding) is not of use to us; however, an "equivalent circuit" will be of value, two iterations of which are given below:

  1. In its simplest form, the internal workings can be viewed as an op-amp which is connected as a voltage follower--its output is tied directly to its inverting input. The non- inverting input (not available on the outside of the chip) goes through a 1.25V "battery" to the Adjust pin; in this way, the voltage follower is displaced from the Adjust pin by 1.25V.
  2. When the above fictitious device is connected in the boot-strapping configuration, it can be seen that the output of the op-amp will pull up on the external resistors until 1.25 volts develops across the "sensing resistor," balancing this against the 1.25V battery in the regulator. An attempt to pull down on the output with a load would, at the same time, decrease the voltage across the external sensing resistor, thus forcing the voltage follower upward as determined by the internal battery. As a result, the output of the op-amp would correct for a drop in voltage.
  3. The above equivalent circuit does not account for the current which exists in the Adjust pin, nor does this circuit depict the true nature of the regulator's "pass transistor." The following is a more complete model.
  4. Instead of a battery, a 1.25V Zener diode is used with its cathode on the non-inverting input and its anode at the "Adjust" pin. Supplying the Zener is a 50uA current source.
  5. The output of the op-amp is no longer the output of the regulator, nor is it tied to the inverting input. Instead, this output runs into a Darlington amplifier; the op-amp output goes to the base of a medium- power NPN transistor, with the emitter of this unit going to the base of an NPN power transistor. tor. The collectors of both transistors go to the unregulated Input. The emitter of the power transistor is the Output and is tied to the op- amp's inverting input (thus including the Darlington amplifier in the nega- tive feedback loop.) (This Darlington amplifier explains why you cannot connect a load so as to pull up on the output of the supply. Current must always be drawn with respect to ground.)

Strictly from an external point of view, the operation of these devices can be restated as follows:

The Output terminal goes through Rl, the "sensing resistor" of perhaps 240 ohms, then through R2, the rheostat, to ground. The junction of Rl and R2 goes to the Adjust pin. No matter where the rheostat is set, the Output will pull on this resistor string until 1.25V is developed across Rl. If the setting of R2 is changed, the Output changes accordingly to re-establish this 1.25V condition. The desired condition occurs when a current of VREF over Rl (1.25V over 240 ohms) is established through this "sensing resistor."

As promised by the manufacturer, putting a load on the Output causes no change in the Output voltage, VOUT. We can assume, then, that a load has no effect on the currents in the boot-strapping circuit. Therefore, the following line of reasoning can be used to derive a formula for the Output voltage as a function of the resistors in the boot-strapping network.

Kirkhoff's Law says that our VOUT will be equal to the sum of the voltages around the resistor string of Rl and R2. (VOUT equals the V across Rl, plus V across R2.) We will consider these two elements separately as follows:

  1. The manufacturer assures us that the voltage across Rl will be maintained at VREF, 1.25V. The resultant current, VREF over Rl, is the main component of the current in R2, and we can use it in the calculations to follow.
  2. There are actually two suppliers of current in R2, VREF over Rl, plus IADJ (the latter being current supplied by the Adjust pin). The voltage across R2 then becomes R2 times the fraction VREF over Rl, plus R2 times IADJ.

Thank you, Mr. Kirkhoff. The grand total is VOUT equals VREF, plus R2 times VREF over Rl, plus R2 times IADJ.

Comparing the two currents through R2, we see that IADJ is only 50uA, whereas VREF over Rl (in our example) is 1.25V over 240 ohms or 5.2mA. IADJ is only about 1 percent of the current through R2; it affects VOUT even less than this. Therefore, we can drop its term in the formula so as to get the expression:

VOUT approximately equals VREF, plus R2 times VREF over Rl.

Or:

VOUT is 1.25 plus R2 times l.25 over Rl

[Zealous arithmetists use the associative property of numbers to obscure the meaning and generate the following formula given in the book: VOUT equals VREF times the quantity 1 plus the fraction R2 over Rl, all the above plus IADJ times R2. This expression is equivalent.]

Specifications

These units have short-circuit protection and thermal shutdown. Their internal voltage standard, VREF, is rated to be within the limits of 1.2V minimum and 1.3V maximum, with 1.25V being typical. The "drop-out" voltage below which these regulators cease to function is about 2 volts above VOUT. The current in the Adjust terminal is typically 50uA, 100uA maximum. Unless the device carries an "H" or "HV" in its suffix, the maximum unregulated voltage allowed between Input and Adjust pins is 40V; units of the "H" persuasion can accept up to 60V.

A wide variety of case styles is available, each of which carries its own restrictions on current handling and power dissipation. The following list explains the suffixes (which denote the case styles) and their ratings:

As an example of how these figures are used, let us consider using an LM3l7T (the popular TO220 package) in a variable supply whose unregulated voltage is 40V. The above table tells us that our regulator is good for l.5 amps, l5 watts--whichever comes first. With the regulator adjusted for 30V output, the pass transistor will have across it a l0V drop; under full load, it will be asked to dissipate l5 watts (l0V times l.5A). However, if the supply were set for l0V output, 30V would appear across the pass transistor; an attempt to draw l.5 amps would be asking the unit to dissipate l.5 times 30, or 45 watts--this is not possible.

Circuits and Design Considerations

In this article, no great emphasis will be placed on the input system--the unregulated supply preceding the regulator. Sample circuits can be found in the original article, "Three-Terminal Monolithic Voltage Regulators," SKTF, Fall l980, and in "A Talking Meter . . .," SKTF, Fall l98l. Another example will be presented in the "Split Supply Circuit" presented in this section. Meanwhile, in the ham radio operator's bag of tricks are the following rules of thumb:

If a center-tapped secondary is used into a full-wave rectifier (using two diodes), the current rating of the transformer should be about 1-1/2 times the desired DC load current. If a full-wave bridge is used across the whole secondary winding, the transformer should be rated for twice the DC load current. The secondary voltage should be chosen so that the "peak" output of the filter (l.4 times the RMS secondary voltage) is a few volts above the "drop-out" point of the regulator.

A general rule in picking filter capacitors is to use 2,000uF per ampere of load current. The "working voltage" of the capacitors should be higher than the voltages which they expect to see.

Basic Adjustable Positive Supply Circuit

A suitable unregulated voltage goes to the Input terminal of the regulator (LM3l7T, for example). To suppress oscillations, this Input is bypassed to ground with 0.luF (located near the regulator). The Output of the regulator goes through Rl (240 ohms), then through R2 (5K rheostat) to ground; the junction of these resistors goes to the Adjust terminal. The supply's output terminals are the regulator's output and ground.

Improved transient response can be obtained with a bypass capacitor from Output to ground. However, if this capacitor is very large, the possibility exists for it to discharge back through the regulator and damage the device. Therefore, the literature suggests that if this unit is larger than 25uF, a protection diode should be connected between the regulator's Input and Output (cathode on the Input).

Improved ripple rejection can be gotten by bypassing the Adjust terminal with l0uF. Once again, a protection diode is required; it is connected across Rl with its cathode on the Output.

Confusion exists throughout the literature about picking values of the boot-strapping resistors. The basic circuit for the 3l7 shows the above values, while others show Rl as l20 ohms and R2 as 2K. The following can be said regarding these choices:

Rl should be chosen so as to draw a certain minimum load current; otherwise the Output will float about aimlessly. The specifications list this required load current as ranging from 3.5mA minimum to 7mA maximum. Given a minimum VREF of l.2V and a maximum required load current of 7mA, it would seem that a safe value for Rl (which would work in all cases) should be l70 ohms, l60 ohms being a standard resistor value. For the sake of argument, all the rest of the circuits will use l20 ohms, as is often done in the Applications Notes.

Picking an appropriate R2 means deciding on what range you want your adjustment to cover. (The upper limit had better be 2 volts below the "ripples" in the unregulated source. The lower limit can be no lower than l.25V, as determined by the regulator.) We can calculate R2 based on the upper limit by using our approximate VOUT equation--rearranging the terms as follows:

VOUT minus VREF equals R2 times VREF over Rl, and therefore:

R2 equals the quantity VOUT minus VREF, all times the fraction Rl over VREF,

Or:

R2 equals the quantity containing the ratio of VOUT over VREF, minus l, this quantity times Rl.

Suppose we want an adjustable supply with an upper limit of 25V. Suppose also that we have chosen an Rl of l20 ohms. Plugging these values into the above, we have: 25 over l.25 (equals 20) minus l, all times l20 ohms. The answer is 2280 ohms, 2.5K being a standard value.

Adjustable Split Supply Circuit

(This is tailor-made for those who wish to experiment with op-amps.) This circuit uses both a 3l7 and a 337, running off complementary full-wave rectifiers to get plus and minus voltages about ground. Each half is designed to operate from l.25 to l5V at one-half amp.

A Stancor 36V lAmp center-tapped transformer is used (Stancor P867l). One side of the primary winding goes to one side of the AC line, while the other side goes through a l/4Amp fuse, then through an on-off switch to the other side of the AC line. The center tap of the secondary is grounded.

For the positive supply, each end of the secondary goes to the anode of a diode (lN4003); the cathodes are tied together and go through l000uF to ground (50V electrolytic, negative at ground).

These cathodes also go to the Input of an LM3l7T--this Input being bypassed to ground by 0.luF. The Output of the 3l7 goes through l20 ohms, then through a l.5K rheostat to ground. The junction of these two resistors goes to the Adjust terminal; this is also bypassed to ground by l0uF (25V electrolytic, negative at ground). A diode (lN400l, 4003, etc.) has its cathode on the Output and its anode on the Adjust pin.

The 3l7 Output, which is the positive output of the supply, is bypassed by anything from luF on up. This Output also goes to the anode of a diode (lN400l, 4003, etc.), with the cathode of this diode going back to the regulator's Input and to the cathodes of the rectifiers.

For the negative supply, each end of the transformer secondary goes to the cathode of a diode (lN4003); the anodes of these diodes go through l000uF to ground (50V electrolytic, positive at ground).

These anodes go to the Input pin of an LM337T--this Input being bypassed to ground by 0.luF. The Output of the 337 goes through l20 ohms, then through a l.5K rheostat to ground. The junction of these resistors goes to the Adjust pin; this terminal is also bypassed to ground by l0uF (25V electrolytic, positive at ground). A diode has its anode on the Output and its cathode on the Adjust pin.

The 337 Output, which is the negative supply output, is bypassed by anything from luF on up (25V electrolytic, positive at ground). This Output goes to the cathode of a diode (lN400l, 4003, etc.), with the anode of this diode going to the 337 Input and to the anodes of the rectifier diodes.

High-Current Adjustable Positive Supply Circuit

This circuit uses a 3l7 in conjunction with three "monolithic power transistors" (LMl95, 295, 395) which are good for over l amp each. [More on these transistors later.] The connection of these transistors is that of parallel emitter followers; their collectors are tied together and go to the unregulated supply, while their emitters are tied together and become the output of the supply. The bases are tied together and go through 500 ohms to the emitters. These bases go to the collector of a PNP transistor (2N2905).

The emitter of the 2N2905 goes to the unregulated supply and to the collectors of the pass transistors. The 2905 base goes through 5.lK to the Input of an LM3l7, with this Input also going back through 22 ohms to the unregulated supply.

The emitters of the pass transistors, which are tied together, go to the Output of the 3l7. This Output also goes through l20 ohms, then through a 5K rheostat to ground. The rheostat is bypassed by l0uF (negative at ground), while the l20 ohm resistor is shunted by a diode (cathode on the output). The output is bypassed by 47uF (negative at ground). This Output also requires a 30mA load--from Output to ground in order to turn on the PNP transistor (forward-biasing it through the 22 ohm Input resistor). At the supply's lower limit of l.25V, this load resistor will have to be 39 ohms.

[Editor's Note: It is unclear to me why they don't affix this 30mA load permanently into the system by changing the l20-ohm sensing resistor to 39 ohms, then changing the rheostat to a l.5K 2-watt unit. It's worth a try, since a 39-ohm Output resistor will have to dissipate 30 watts when the supply is turned up. Also in their circuit, the rheostat is generous by about 35 percent; as it stands, there will be a dead spot over the top third of its range.]

The pass transistors (LMl95, LM295, LM395) are not simple transistors, but are monolithic IC's. A figure for "Beta" is not meaningful, since their base current is a relatively constant 3uA over a wide range of base-emitter voltages; they are perhaps best described as voltage-controlled, high-current devices. As stated in the literature, they lend themselves to being paralleled such that one could use this same circuit to control a hatful of them (the additional "base current" of added units would not be significant).

Whether l95, 295, or 395, they are all rated at "over l amp"; the l95 and 295 are good for 42V, while the 395 is good for only 36V. (The principal difference between these three is in temperature specifications, the l95 being the most tolerant of hot and cold.) LM395's with the "T" suffix are in a TO220 package and cost about $2.50 each. Units with the "K" suffix are in a TO3 can, and cost about $6.50 each. It seems that their cases are not common to the collector, as is usual; the case is the emitter in these units.

Regulation of High Voltages

As long as the Input-to-Adjust voltage is restricted to 40 volts, these devices can be used at any voltage above ground. The previously described circuits can be used by lifting the grounded end of the rheostat and putting it atop a high-voltage Zener diode or gas VR tube. However, the 40V functional range of these devices is something to keep in mind; this imposes severe restrictions at voltages where 40V is not a significant percentage. At 400V, for example, a 40V restriction on the Input range represents a total of l0 percent--only variations of plus or minus 5 percent can be tolerated in the unregulated supply.

The literature contains a circuit for a l00V adjustable supply at unspecified current which uses an LM3l7 in conjunction with an unspecified triode vacuum tube.

Adjustable l00V Positive Supply Circuit

The unregulated input (unspecified) goes to the plate of a triode tube; its cathode goes to the Input of a 3l7. The Output of the 3l7 (which is also the output of the supply) goes to the grid of the tube. From Input to Output of the 3l7 is a 40V Zener diode, its cathode on the Input. From the Output of the 3l7 to its Adjust pin is 600 ohms; a 50K rheostat goes from Adjust to ground. [One wonders what happened to the minimum load restriction on the 3l7, in addition to which, 600 ohms is hardly a standard resistor value.]

The Input of the 3l7 is bypassed to ground by 0.luF. The Output is bypassed to ground by luF (200V electrolytic, negative at ground).

Of course, the insulating materials used in mounting the regulator must withstand the voltages being considered. Capacitors at appropriate voltage ratings must also be chosen for use around these circuits.

Pin Connections

National Semiconductor Stereo Power Amplifier IC's

Abstract

A survey of stereo amplifier IC's capable of delivering 2, 4, and 6 watts per channel is presented. Example circuits are given, including one with a transistor final output stage, one having a "tone control," and monaural amplifiers using both halves of the chip in combination to get double the power. Their uses would include building a portable power amplifier and loudspeaker system for "Walkman-type" equipment, building a monitor amplifier for tape decks being used in on-site recording, or perhaps for supplying sufficient power in their monaural connection to drive low-efficiency tactile transducers.

Introduction

The devices discussed here are "power op-amps" capable of delivering generously high currents to loudspeakers (primarily to 8- or l6-ohm loads). An overall survey of several IC's imposes practical limits on our discussion as follows:

Fine points such as individual distortion characteristics will not be tabulated here; it is hoped that general summary statements will suffice. Likewise, the issue of picking the smallest heat sink to serve the designer's need for power dissipation will not be tabulated; it is hoped that a couple of examples will suffice. Finally, these beefy op-amps have applications other than audio--for example, the LM379 is shown in its Applications Note as a driver for 2-phase motors. We shall not fill your bookshelves with all this material, but appropriate literature will be listed with the pin diagrams at the end of this article. The following chips will be discussed in this survey:

  1. LM377, Radio Shack 276-702--Two watts per channel in a l4-pin DIP (although supplanted by the improved LMl877, it is included because of its availability from Radio Shack).
  2. LMl877--Two watts per channel in a l4-pin DIP (a pin-for-pin replacement for the above 377, it has superior distortion characteristics).
  3. LM378--Four watts per channel in a l4-pin DIP (very similar to the above, this chip can operate on higher supply voltage).
  4. LM379--Six watts per channel in a l4-pin DIP (this package carries a partial heat sink which can be bolted down to a larger sink).
  5. LM2877--Four watts per channel in an ll-pin, single in-line package (has a mounting tab for attachment to external heat sink).

Summary of Specifications

All units have current limiting and thermal shutdown. Other than the 2877, which is shown as driving a 4-ohm load, the other units are recommended for 8- or l6-ohm loads. All units require heat sinks if their rated power is to be realized; the l4-pin packages have multiple grounding pins for connection to a heat sink, while the 379 and 2877 have provisions for being bolted to heat sinks.

So as not to require a dual (split) power supply (a plus and a minus supply as seen in traditional op-amp circuits), these chips contain a built-in voltage divider (made up of two l5K resistors across the single VCC supply). The output of this divider is a VREF of l/2 VCC, available at a "bias pin." (This pin is not present on the outside of the familiar LM386 amplifier, but it is very much a part of its inner workings.)

At medium power levels and at low frequencies, the total harmonic distortion (THD) is less than 0.2% for all the units. Also in every case, this distortion goes up markedly (in some cases exceeding l percent) above l0kHz; who cares, we can't hear the second harmonic of l0kHz.

At higher power, these devices are not so impressive. The 379 gives you a THD of nearly 3% as the output approaches clipping. The 2877 has a THD of l0% when it delivers its 4 watts into a 4-ohm load.

The old 377 and the high-powered units have "cross-over" distortion which becomes more significant at low power levels. (This is reminiscent of cross-over distortion in early transistor hi-fi amplifiers whose output transistors were biased near cut-off at quiescence.) The distortion is a bit higher than 0.2%, even at low frequencies, when running at a power level of 0.05 watts.

Facts About Power

First of all, the realizable power is limited by the usable voltage swing of the amplifier's output; this limit being imposed by the supply voltage which runs the chip (as well as current limiting which will occur if the load impedance is low). For example, two watts of power can be delivered to an 8-ohm load only if an RMS voltage of 4V can be imposed across it. The power "P," the voltage "E," and the load resistance "R" are related by:

P = E2/R

Plugging in an R of 8 ohms and a power of 2 watts, we have: 2 = E2/8 -- E2 = 2(8) or l6 -- E = 4V.

Using a sine wave for our analysis, we can define the necessary voltage swing as follows: The peak value of a sinusoid is l.4l4 times the RMS voltage; this gives us the distance which the output must swing in either direction from quiescence in order to give us our needed 4V RMS signal. Peak swings of equal amplitude in either direction must be made, thus requiring that the output be functional over a so-called "peak-to-peak" voltage range of 2 times l.4l4 times our 4V RMS value, or ll.3V.

The op-amps in these packages can only bring their outputs to within about 2.5 volts of the power supply extremes--from 2.5V short of VCC down to 2.5V above ground. In order to get a good, comfortable peak-to-peak swing of ll.3V, the supply must be 5 or 6 volts higher than this, perhaps l7 or l8 volts. Therefore, in order to get 2 watts per channel from any of these devices, an l8V supply is required.

From the above, it can be seen that we won't get full power by running these chips from a l2-volt battery. Just for fun, we'll perform the above calculations backwards to see just watt can be realized.

From a l2V supply, we must throw away at least 5 volts to allow for the boundaries of the output range, thus giving us a functional peak-to-peak range of 7V. Our attainable RMS output voltage is gotten by dividing 7V by 2.828 (twice l.4l4), giving us 2.475V RMS. "P" equals 2.475 squared over 8, which is only 3/4 watt per channel.

It is interesting to note that the main discernible difference between the 4-watt and 2-watt chips (the 378 vs. the 377 and l877) is in maximum supply voltage ratings; the 378 has a rated maximum of 35V, while the low-power units specify a maximum of 26V. Their rated "package dissipation" is the same, 4 watts.

Heat sinks are essential if these devices are expected to operate at full power. While no damage will occur if the heat sink is insufficient, partial thermal shutdown tends to cause gross "cross-over distortion" and "flat-topping" as the chip temperature rises. These problems disappear as the chip is cooled.

With a small heat sink, such as a 2-1/2 inch square of foil on the top side of the PC board, the dissipation of heat will seriously limit the audio power output as the supply voltage is increased. For example, using a 377, the maximum power output which you can expect from such a system will be l.l watts per channel at a VCC of l8V; this will decrease to 0.l5 watts per channel at a VCC of 22V. Attaching an external heat sink to this same chip will solve this problem to the extent that the power output will increase with a higher VCC.

Maximum power for any given setup will be gotten when the choice of VCC (so as to define an available voltage swing) is balanced against the heat sink's ability to handle the resultant power dissipation (4 watts in the best of cases for the smaller units). (Notes for the smaller units recommend a specific piggyback heat sink, a Staver V7-l, but they also admit that this system will not meet the 4-watt dissipation spec of the ideal situation.) In short, the rated power of these devices is available as advertised, only if you run them off of Hoover Dam and listen to your stereo in the icehouse.

On the other hand, you need far less power than you might expect to get a good listening volume, especially from efficient loudspeakers. For example, I wired up the monaural "bridge" amplifier described here, using a 377 and a l2-volt power supply. (As a heat sink, I mounted a piece of copper-clad Vector board atop the chip.) The resultant system, amounting to slightly over l/2 watt, had a microphone on its input and a l2-inch TV speaker on its output. As soon as I did my Bing Crosby imitation, the degree to which nearby office doors were slamming was impressive.

By the way, loudspeaker efficiency as a function of size is badly misunderstood. For any given baffle design, "air suspension," "bass reflex," or "infinite baffle," the efficiency goes up as the speaker gets larger. Those who say "Well, I don't need very large speakers because my amplifier is quite small," have their logic backwards; the bigger the diaphragm, the more air it can move. This does not apply when switching to a different type of baffle; the little speaker with the pseudo "infinite baffle" in my transistor radio is more efficient than someone's "air suspension" Wharfedales, that's true.

Practical Building Considerations

The 377, the l877, and the 378 have 6 grounding pins for connection to a heat sink. Outboard units, such as the Staver V7-l heat sink,* are recommended in the literature, although these units will not dissipate sufficient heat to bring you up to full power output. I recommend the following schemes for the home builder.

*Staver Company, Inc., 47 North Saxon Ave., Bay Shore, NY ll706.

Perforated Vector Board can be gotten which is clad with copper on one side (such as the l69P44Cl, 4-l/2 by l7 inches). Vector also sells a tool which cuts through the foil so as to isolate individual holes from the rest of this "ground plane" (Pl38A). Point-to-point wiring can be done as usual on the non-foil side, using this tool to isolate each and every hole receiving an ungrounded component lead. Before a component is inserted, the tool is used to isolate this hole from the foil; this should be done for the active pins of the chip as well. The grounding pins of the chip can then be soldered to the foil on the top (component) side.

In testing a couple of these circuits, I used a piece of copper-clad Vector Board which I attached to the top of the chip. I bent 5 of the 6 grounding pins backward so as to be sticking straight upward. Then a piece of the copper-clad board (foil side up) was set onto these pins and soldered.

A small copper plate could be used as a heat sink by placing the chip on its back and bending 5 of its 6 grounding pins down against the plate. When soldering these pins to the plate, you would do well to insulate the top of the chip from the hot copper with a little slip of braille paper. Also, soldering will go a lot faster if you pre-tin the plate.

Remember which grounding pin you leave sticking down to fit into the socket or PC board; with a heat sink mounted atop the chip, no other indication will be present as to the location of pin l.

The 379 carries with it a partial heat sink which extends beyond the ends of the chip, bending down so as to meet the mounting surface plane. Tapped holes in these end flanges permit bolting of the unit to a heat sink plate.

The 2877 is a long package, the upper edge of which has a tab with a mounting hole by which it can be secured to an external heat sink.

Typical Circuits

The following circuits are taken from the National Semiconductor literature; they are typically shown for all of the chips. Noting the similarity among circuits for these various chip numbers, it is obvious that the main differences between chips are their packaging and/or maximum supply voltage ratings.

Only one channel of each case is given here, since connections for both are identical. In all cases, the grounding pins are tied together and go to the negative side of the supply. The VCC pin goes through a switch to the positive supply. Except where this supply is regulated (and to some extent, even then), the VCC pin should be bypassed with a healthy capacitor of 500uF or greater (rated at greater than VCC and with its negative end grounded). Also in every case, the bias pin is bypassed with a capacitor whose rated voltage must be greater than l/2 VCC (negative at ground). (Values of the latter unit will be given individually in the circuits.)

Non-Inverting Amplifier

As we know from "Operational Amplifiers," Winter l983, non-inverting amplifiers have a characteristically high input impedance (in this case, l00K as set by a resistor). If the bias pin were low-impedance, one could return the inverting input's resistor directly to it, thereby leaving the non-inverting input to serve as an "infinite-impedance" input. This is not possible here; the bias pin is taken from a voltage divider inside the chip, which is made up of two l5K resistors, hence dictating the bias arrangement shown. Aside from this exception, this non-inverting amplifier is a textbook case.

Non-Inverting Circuit

The cold side of the input signal is grounded. The hot signal lead goes through 0.luF to the non-inverting input of the op-amp. This non-inverting input also goes through l00K to the bias pin, with this bias pin being bypassed to ground by 250uF.

There is a feedback resistor of l00K going from output to inverting input. This inverting input also goes through an "input resistor," then through 5uF to ground (good for l/2 VCC, negative at ground). Two values of "input resistor" are shown; 2K gives you a gain of 5l (l plus l00K over 2K), while the other is 5l0 ohms, giving you a gain of about 200 (l plus l00K over 0.5l0K equals l97).

The output goes to the positive end of a coupling capacitor (200uF or greater, rated at greater than VCC). The negative end of this capacitor goes through the speaker to ground.

Inverting Amplifier

Without exception, this circuit is straight out of the bastions of Academe. It has one major disadvantage; its input impedance is that of its input resistor--this impedance gets low as the gain is increased. Two circuits are given, one whose gain is unity and whose input impedance is reasonable, and the other whose gain is 50, having an input impedance of 2K. As we know, the gain is the feedback resistor divided by the input resistor.

Unity Gain Inverting Circuit

The bias point, which is bypassed to ground by l0uF, goes to the non-inverting input. The output goes through l00K to the inverting input, with this inverting input going through l00K, then through 0.luF to the hot input signal lead. The cold side of the input signal is grounded. As in the non-inverting circuit, the output goes through a large coupling capacitor, then through the speaker to ground.

High-Gain Low-Impedance Inverting Circuit

The bias pin, which is bypassed to ground by l0uF, goes to the non-inverting input. The output goes through a l00K feedback resistor to the inverting input. This inverting input goes through 2K to the positive end of a 5uF input coupling capacitor (rated above l/2 VCC), with the negative end going to the hot signal lead. The cold side of the input signal is grounded.

As before, the output goes to the positive end of a large coupling capacitor (200uF or greater, rated above VCC), with the negative end of this capacitor going through the speaker to ground.

Stereo Amplifier with "Bass Tone Control" for Ceramic Phono Pickups

Set for a flat response, this amplifier has a bass roll-off of about 75Hz. With a cross-over point at about lkHz, the control can be used to cut the bass l5dB and to boost it slightly more than l5dB (this measurement being done at l00Hz).

The cold side of the ceramic cartridge is grounded. The hot side goes to the top of a lmeg audio-taper volume control, the bottom of which is grounded. The arm of this control goes through 0.luF to the non-inverting input; this non-inverting input also goes through lmeg to the bias pin, which is bypassed by 50uF.

The inverting input goes through 5lK, then through l00uF to ground (negative at ground, rated at greater than l/2 VCC). This inverting input also goes through a 5l0K feedback resistor to the arm of a l00K pot (bass control). The top of this pot goes through l0K to the output of the op-amp; between the arm and the top of the pot is connected 0.033uF. The bottom of this pot goes through lK to the junction of the 5lK input resistor and the l00uF capacitor; between the arm and the bottom of the pot is connected a 0.33uF capacitor.

The output goes to the positive end of a very healthy coupling capacitor (250 or 500uF, rated at greater than VCC), the negative end of which goes through the speaker to ground.

Inclusion of an Add-On Transistor Amplifier

Listed as a "l0-watt amplifier," this circuit can be added to any of the chips. However, given the supply constraints as discussed above, as well as recognizing that an additional voltage will be lost across the transistors, the only time you would get to see this "l0 watts" is into a low-impedance load (4 or 8 ohms). There's no way around it, the supply voltage must be as high as you can muster for this setup to be advantageous.

The circuit uses a "complementary pair" of power transistors to make the op-amp system beefy indeed. As can be seen, they are included in the feedback network, and thus they become part of the operational amplifier. (A complementary pair is made up of an NPN and a PNP emitter follower connected back-to-back.)

Circuit with Transistor Amplifier

The cold side of the input is grounded. The hot input lead goes through 0.luF to the non-inverting input. This non-inverting input goes through l00K to the bias pin; this bias pin is by-passed to ground by 250uF (negative at ground). The inverting input goes through 2K, then through 5uF to ground (negative at ground).

The collector of a PNP transistor is grounded; the collector of an NPN transistor goes to VCC. (The PNP unit is a 2N5l94 or National NSP5l94. The NPN is a 2N5l9l or National NSP5l9l.) The VCC line is bypassed by l000uF (negative at ground, and rated at greater than VCC).

The two bases are tied together and go to the output of the chip. These bases also go through 5 ohms to the emitters, which are tied together. These emitters are the output of the system; they go through a l00K feedback resistor to the inverting input on the chip. To suppress oscillations, this feedback resistor is shunted by 82pF in series with 27K. The emitters also go to the positive side of a l000uF capacitor, the negative side of which goes through the speaker to ground.

Monaural "Bridge" Amplifier

The two op-amps in the package can be connected in opposition to get a monaural amplifier of twice the power. This configuration is called a "bridge" amplifier. The speaker is connected between the two op-amp outputs; any jack for the speaker or earphones must be appropriately insulated from ground.

One op-amp of the chip is set up as a non-inverting amplifier, with the other being connected as an inverting amplifier. The non-inverting section has the input signal capacitively coupled directly into its non-inverting input. As was discussed in "Op-Amps," Winter l983, an op-amp in the non-inverting connection has a replica of the input information on its inverting input--a "virtual input signal." It is this virtual input signal that drives the other half of the chip, an inverting amplifier which necessarily has a low impedance.

Another trick used in this circuit is that both op-amps share a common input resistor--in this example, l0K. (A coupling capacitor is placed in series with this resistor to eliminate offset effects from direct coupling. For the purpose of analysis, let us pretend that this 0.47uF capacitor is not there.)

With respect to the inverting amplifier, everything looks straightforward and standard. Its non-inverting input is fixed to a bias point, VREF. It has a feedback resistor of l megOhm going from its output back to its inverting input. Its inverting input goes through a l0K input resistor to a signal source (in this case, a "virtual signal point" on the other op-amp).

With respect to the non-inverting amplifier, an external signal is applied to its non-inverting input. Its gain-determining network consists of a l megOhm feedback resistor (from output to inverting input) and a l0K resistor from its inverting input to a fixed "common" (in this case, "common" is a "virtual bias point" on the other op-amp).

The l0K resistor mentioned in both cases is the same unit, common to both sections. The gain of each stage is calculated in the usual way; for the inverting amplifier, the gain is l meg over l0K equals l00, and for the non-inverting amplifier, the gain is l plus the ratio of l meg over l0K equals l0l. The speaker, straddling the outputs like a wishbone, sees both gains--the resultant bridge amplifier has a gain of 20l.

Bridge Amplifier Circuit with Gain of 20l

The grounding pins on the chip are tied together and go to the negative side of the power supply. The positive side of the supply goes through a switch to the VCC line, this line being bypassed to ground by l000 or 2000uF (negative at ground). The VCC pin on the chip goes to the VCC line. The bias pin on the chip (VREF) is bypassed to ground by 250uF (negative at ground).

The input signal (either from the hot side of an input jack or from the arm of a volume control) goes through a Mylar 0.luF capacitor to the non-inverting input of A1. This non-inverting input also goes through l00K to the bias pin. There is a lmeg feedback resistor from output to inverting input on A1.

The non-inverting input of A2 goes to the bias pin. The output of A2 goes through a lmeg feedback resistor to its inverting input. The A2 inverting input goes through 0.47uF (Mylar), then through l0K to the inverting input of A1. The speaker is connected between the two outputs.

Pin Connections

(Note: Catering to technicians who are not, as we are, students of op-amps, the manufacturer has chosen more simplistic labels for the input pins--"Input" for the non-inverting input and "Feedback" for the inverting input. We sophisticates will get more use out of these devices if we call their pins by their true colors.)

LM377, LMl877, LM378, LM379:

(These and other circuits are given in National's Applications Note AN-l25.)

LM2877:

(Circuits discussed in National's book, "The Audio/Radio Handbook, l980.)

On the above single in-line package, the pins are labelled l through ll, from left to right. The pin 1 end of the package is marked in two ways; there's a small cutout on the end adjacent to pin 1, and a diagonal cut on the corner of the mounting tab is larger than at the end of pin 11.

[Editor's Note: I have just been informed that the brand new Radio Shack listing for the 276-702 is LMl877, the improved version. However, this change is so recent that stores with old stock may still sell you a 377 under that same number.]

The Editor's Evaluation Of The DRI Industries' One-Hand Soldering Iron

Abstract

This discussion of the DRI Industries'* "One-Hand Soldering Shop" iron is in two parts. The first is an honest attempt to objectively assess the merits of the device as a tool. The second outlines reasons why the Editor feels that thumbwheel solder-feeding systems are not advantageous to the blind user; these points are questions for debate, and rebuttals for publication here are encouraged.

*DRI Industries, Inc., lll00 Hampshire Avenue South, Bloomington, MN 55438

Part I
Features of the Tool

Inventory

The "Soldering Shop" comes with a rather complete collection of worthwhile accessories, which are listed below:

  1. A well-organized carrying case. (A piece of the foam packing material is intended for insertion into the lid, so do not discard it offhandedly.)
  2. The iron itself, all loaded with solder and ready to go. (A large 1/4-inch wide, chisel-shaped tip comes installed.)
  3. A conical tip to replace the large one mentioned above (not quite small enough for soldering IC's).
  4. A small sheet metal cradle to serve as a rest stand.
  5. Three extra 50-inch rolls of 0.045 inch rosin-core solder, along with two extra feeder spools.
  6. Three "soldering aid" probes, a Phillips screwdriver (for performing surgery on the iron), a clip-on heat sink, a pair of self-closing tweezers, and a putty knife.
  7. A can of "Nocorode" zinc chloride flux paste (not for electrical work), and a small brush for applying it. (Residue from this flux should be rinsed away with water.)
  8. Assorted short lengths of "heat shrinkable" tubing.
  9. A packet of assorted hardware--everything from large cotter keys to wood screws.
  10. Brochures on other DRI Industries' products, primarily cabinets, work benches, and hardware assortments.

Description

The device is a 60-watt iron of the "constant-power" type (not temperature controlled). It has a pistol grip, the upper left corner of which has a thumbwheel for feeding solder to the work. The bottom end of the handle contains a spool of solder, the free end of which is threaded up through the handle; it emerges from a tube which is underneath and parallel to the heating element. The tip is bent downward so as to meet the solder about 5/8 of an inch beyond the forward end of the feeder tube.

The unit takes 2-l/2 minutes to heat up to soldering temperature. It remains hot enough to melt solder up to 5 minutes after being unplugged, and it should not be touched for more than l0 minutes after being disconnected.

I do not view the unit's relatively high power (60 watts) and its high heat capacity (thus the long cooling time) as faults of the tool. Those of you who have read "Soldering," Parts I and II, know why irons of high power and high heat capacity promote quick action in soldering and do less damage than underpowered units.

By luck or by design, the solder spool in the handle is insulated from the feeder tube, and the feeder tube is attached by a wire clip to the heating element. A continuity tester can be connected between the solder as it emerges from the spool and the feeder tube; this tester will then beep to indicate that the solder has reached the tip. (A spade lug can be mounted under one of the screws which holds the heating element to the handle; this makes a very sturdy permanent connection to the feeder tube. The other tester lead must be attached to the solder via a clip lead.)

Unfavorable Characteristics

The biggest strike against this instrument is its long barrel (heating element), and the fact that set screws for the tip protrude significantly from either side. The long barrel puts the tip 6 inches beyond the user's hand, thus greatly increasing the error in reaching the target and magnifying the effects of tremor in his hand. The basic diameter of the barrel, l/2 inch, is not unreasonable, but set screws securing the tip (which also support the forward end of the feeder tube) protrude to make the width of this assembly 7/8 of an inch. The resultant size and shape of the iron is something to be reckoned with --it is relatively commonplace that protrusions and/or the feeder system snag on wiring and holding devices. Inadvertent contact with the user's free hand is also hard to avoid.

The system used for securing the tip is not one to ensure intimate contact between the tip and the heating element; this intimate contact is desired for efficient heat transfer. Ideally, where a set screw system is used, the screws should be placed so as to force the shank of the tip to one side. In this unit, however, the two set screws are directly opposing each other; they hold the tip firmly between them, but serve as a pivot about which the tip can rock up and down in its chuck.

Neither the system holding the tip nor the mounting of the chuck to the heating element is rigid. Up and down motion of the tip can always be accomplished, even accounting for expansion of these components when the iron is heated. (As received, 3 of the 4 mounting screws holding the barrel to the handle were very loose, allowing even more freedom of movement. This is something to check for before turning it on.)

The inherent mobility of the tip with respect to the handle leads to instability in holding the iron against the work, as well as raising questions as to how well its heating efficiency can be maintained (oxides may eventually develop between the various components of the device).

All the above factors, as well as the available tip sizes, define the size of work for which this iron is appropriate. Its use would be difficult for anything smaller than terminal lugs and sparsely populated PC boards.

If I may be permitted to briefly discard my cloak of scientifism, I would like to say that aside from the issue of self-contained solder feeders, this iron is ill-suited for a blind user. I find it very hard to hit a target with the instrument, and they could hear my yelps clear out to the Islands of Langerhans every time I inadvertently brought a knuckle up against that barrel assembly.

Part II
Thumbwheel Solder Feeders and the Blind User

I have personal experience with three such devices--a universal attachment (of unknown make, made perhaps two decades ago) for any soldering iron, a "Free-Hand Solder Feeder" designed for the standard Weller soldering gun, and the DRI Industries device described here. I have not tried the Wahl "Isotip Cordless Iron" (see Winter l98l), which is no longer being manufactured.

The devices I have tried are generally of sound mechanical design and do the job as advertised; they permit the soldering process to be executed with one hand while work pieces are supported with the other. My quarrel with them as a blind user has been that they indirectly address the true problem of how to place the iron on the connection, at the same time stripping away several key bits of information which a blind person uses during the soldering process. In addition, they lead to instability of the hand when their thumbwheel is operated.

If everything were to go right, one could do as the instructions say--hold the iron against the work long enough for it to heat up, then use the thumbwheel to feed solder to the work. There are several variables here which the blind person must account for.

Heating time at the connection depends on several factors. One is the physical size (heat capacity) of the work pieces. Another is the effectiveness with which the iron is held against them. Another is the presence or absence of oxides on the materials and on the tip of the iron. The only way to assess the heating process is to monitor the temperature of the work pieces with the free hand (touching a component lead or the like); as long as your "free" hand is already down there, it might as well have solder in it, yes?

Waiting too short a time before feeding the solder will often result in the solder being used up in cleaning and tinning the tip. Sometimes the unmelted solder will be deflected off to one side by the work pieces, thus no longer finding itself in contact with the materials. Waiting too long before feeding allows rapid oxide buildup at the connection, which prevents soldering.

Because of the solder's proximity to the tip of the iron, initial melting occurs more often than not. However, this does not assure that all components of the work will experience "wetting"; the solder is never fed to the work pieces alone, at a point not in contact with the iron. Localized "wetting" may occur on only one of the components of the work. This, by the way, is quite apparent visually, but is not so obvious from tactile feedback through the iron.

Flux-core solder tends to kink and wrinkle as it is rolled and unrolled; slight irregularities in it often cause binding in the feeder system. Furthermore, since solder is a soft material, one skid of the thumbwheel against its surface creates a "dead spot" from which no motion can be initiated. When this happens, the solder must be pulled forward slightly from the forward end of the feeder tube so as to re-engage with the wheel. None of this is apparent through tactile feedback from the thumbwheel; the wheel turns stiffly whether it is feeding the solder or spinning against it. Therefore, you can never be absolutely sure that you have fed solder to the work, and the amount you feed is always open to question.

The final flaw in all of these devices is that you must move your thumb or a finger to operate the feeder. This causes muscles in the arm to pull tendons through your wrist, a very destabilizing action. It happens frequently to the sighted user that the tool slips off the connection--something for which he says "Oops," and puts it back into position. Re-establishing contact with the target is not so trivial for the blind technician.

I stand by my discussion of "Tactile Feedback" in "Soldering," Part II, Winter l98l, wherein the iron and the solder become "canes" in each hand. With this two-handed system, you can find out where the iron is with the solder, you can tell which items are hot enough to melt the solder, and you rarely overheat and oxidize a connection before initial melting occurs.

Yes, placing the iron on the target is a problem, but I stand by my discussion of "Landmarking," also in Part II, as a viable alternative. For those instances where a probe is needed to guide the iron down to the work, I urge you to try the JA3TBW Solder Guide as another alternative.

It would indeed be wonderful to have an iron, all the operations of which are done by one hand. This is desirable because of the many instances where holding the work pieces in position is necessary. However, although I am philosophically slow to admit it, tools with a thumbwheel feeder system are ones which are more accessible to the sighted user than to the blind.

The JA3TBW Solder Guide

As mentioned in our "Hints and Kinks," SKTF, Spring l982, Mike Bhagwandas (JA3TBW) of Kobe, Japan, invented this guide. Since that time, our lab has done considerable experimenting with its use as an aid to soldering short-legged components which have poor "landmarking" features into printed boards. Blind subjects who have built our prototype "kit sets" have found this instrument indispensable when soldering the IC sockets and trim pot. So that you can enjoy the fruits of this research, we have attached the described version to each copy of the magazine; please wear it in good health.

The material we settled on is stainless steel, and we did so for the following reasons:

  1. It will never char and contaminate the iron.
  2. If "acid" or "stainless steel" flux is not used, it is unlikely that solder will adhere to the tube.
  3. Stainless steel has poor heat conductivity.
  4. The resultant device can be of small diameter; it is available having a wall thickness of 0.009 inches.
  5. This tubing is commonly available, sometimes being known as "hypodermic tubing."

Specifically, we purchased "Hypoflex" seamless thin-walled stainless steel tubing of Grade 304; its dimensions are 0.072 inches outside diameter (o.d.), 0.009 inch wall thickness, giving us what they call a "theoretical inside diameter" (t.i.d.) of 0.054 inches. Unfortunately, this product only comes in bundles of l20 feet at 62 cents per foot. However, a similar product can be gotten from "Small Parts,"* in 6, l2, and 24 inch lengths. They have two types available: HTX-l5 Hypodermic Tubing with a 0.009 inch wall, and HTX-l5TW "Thin Wall" tubing with the same outside diameter of 0.072 inches and a wall thickness of 0.006 inches. The former is $2 per foot; the latter is $l.20.

The size was chosen so as to accommodate solder of approximately 0.03 inches; the tube has plenty of room to spare for wrinkles in the solder to pass freely through it.

Two brands of solder can readily be gotten with these dimensions. The first is Kester, 604A33-lSN6066.03l44, available from Marshall Industries.* This Kester solder, which has a diameter of 0.03l inches, is also available from Mouser Electronics* under their number 533-24-6040-27. Finally, 0.028 inch "Multicore" solder is available as Ersin number SN60 22SWG. The Kester is preferable, since "Multicore" solder tends to contain an overabundance of flux, thus causing gumminess and stickiness in the tube.

The tubing has such a thin wall as to afford being "parted" with a file, especially knife-shaped or three-cornered files. After sectioning off a piece (4 inches being a popular length in our trials), square off the ends with a file; then de-burr the inner edges with the tip of a file or with a drill bit. (We have done all of this to the 4-inch piece you've received.)

For unclogging these units when solder gets lodged inside, a No. 55 drill bit is a handy addition. This bit is 0.050 inches in diameter, and will comfortably ream it out without galling.

One final note on making your own customized version: You may wish to devise a handle so as to afford better control. We would be very interested in the schemes you come up with along this line. A small "shaft stop" with a set screw is included with your guide. Used as an adjustable collar, the shaft stop can be secured to the tube just below your hand so as to promote stability in holding the tube against the work. (The shaft stop is available from Player Piano Company.* It has an inside dimension of 0.098 inches, listed as their catalog number 889 at 20 cents each.)

In operation, solder is passed through the tube so as to just make itself known at the business end, this end being rested on the connection. Solder can be fed to the work with the thumb and forefinger, while some arrangement of the remaining three fingers is chosen to support the tube. Good stability will be promoted if the connection has a feature against which the inner edge of the tube can be braced. For example, I bend socket pins outward once they have passed through the board; I actually set the tube over the intended pin, at which point I lean the tube out to the side so as to permit the iron to reach the pin.

Once you are in position, find the lower half of the tube with the iron and slide it down onto the connection, all the while pressing the solder down against the work. I find this technique to be so easy that glee overtakes me, giving me a tendency to apply too much solder. Remember in using this technique that all the solder will be applied in the right place, for a change, and very little of it will be used up exploring the iron.

If left stationary in the tube after the connection is made, the solder will seize up in the bottom end. This is not due to actual bonding with the stainless steel, but is due to the flux solidifying, acting to glue the solder in place. Reheating the end of the tube with the iron will often loosen things up; otherwise your No. 55 drill can be used to ream the tube clear.

The way to prevent adhesion is to create relative motion between the tube and the solder while the flux is cooling. Some of us pull the tube away from the connection and run the solder through so as to protrude l/4 inch or so. Others withdraw the solder into the tube about l/2 inch or so. I spin the tube while it is cooling. All of this is done after the connection has been soldered; you have plenty of time because the flux must cool considerably before it solidifies.

Like anything, this takes a little practice; you will at first, by applying far too much solder or by heating up adjacent connections with the iron, inevitably bridge pins here and there. However, comparing my success with and without this instrument would be ludicrous--there is no comparison. Thanks to Mike, JA3TBW, there is a way to take the heat off our target practice when the going gets tough.

* Address List of Suppliers